# Kiến trúc phần mềm Radio P8

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## Kiến trúc phần mềm Radio P8

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RF/IF Conversion Segment Tradeoffs This chapter introduces the system-level design tradeoffs of the RF conversion segment. Software radios require wideband RF/IF conversion, large dynamic range, and programmable analog signal processing parameters. In addition, a high-quality SDR architecture includes specific measures to mitigate the interference readily generated by SDR operation. I. RF CONVERSION ARCHITECTURES The RF conversion segment of the canonical software radio is illustrated in Figure 8-1. The antenna segment may provide a single element for both transmission and reception....

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4. 268 RF/IF CONVERSION SEGMENT TRADEOFFS Figure 8-2 Superheterodyne receiver architecture. Figure 8-3 Frequency plan suppresses spectral artifacts. Artifacts must be controlled in the conversion process [241, 242]. In addi- tion to the desired sideband, the conversion process introduces thermal noise, undesired sidebands, and LO leakage into the IF signal as shown in Figure 8-3. Thermal noise is shaped by the cascade of bandpass and low-pass filters. Depending on the RF background environment, thermal noise in the receiver may dominate or thermal-like noise or interference from the environment may dominate the noise power. Superconducting IF filters suppress noiselike interference generated in one cellular half-band from a second, immediately adjacent half-band (e.g.,
7. RECEIVER ARCHITECTURES 271 complicated handset design. Recently, however, CMOS silicon RF 50 W to 40 GHz has been reported. One of the .18 micron CMOS chips [250] supports 2.4 GHz RF at 1.8 Volts. CMOS devices that have been demonstrated include low-noise amplifiers, mixers, differential oscillators, IF strips, and RF power amplifiers with 1 W output and 40 to 50% efficiency at 1 to 2 GHz [251]. C. Digital-RF Receivers PhillipsVision [252] created some excitement by announcing a software-radio on a chip. The interesting aspect of their product announcement is that the de- modulator is said to “operate at RF.” Due to the necessarily vague nature of the statements, it is impossible to determine the exact nature of the demodulation process. This announcement plus the recent interest in digital demodulation at RF makes it useful to address this alternative. The comments below may not be representative of the PhillipsVision product, but they reflect research approaches to digital demodulation at RF. Since GHz clocks can be fabricated in single ASICs, one may employ such a clock to demodulate certain modulation types at RF. One approach is the one-bit direct conversion digital receiver, which may be called the RF zero-crossing demodulator. With this approach, the RF is amplified until it is hard-limited into a square wave. Reference square waves are synthesized for each channel-symbol state. An MSK waveform, for example, has two square waves. One corresponds to the mark, say, the lower of the two frequencies. By generating digital streams at mark and space frequencies and counting the number of coincidences between mark and space streams in the incoming RF signal, one can estimate the state of the RF waveform. A bit-timing logic state- machine can then determine bit timing to produce the baseband bitstream. All this can be implemented for a single channel-modulation type in an FPGA using less than ten thousand gates. One advantage of this architecture is that the bit patterns for the channel states may be stored in a lookup table. Differ- ent waveforms at different frequencies correspond to different lookup tables. By using clever data-compression techniques, the lookup tables may be kept compact in spite of the large number of entries in the table. A similar approach simply counts zero-crossings of the RF. Once the vari- ance of N zero-crossing counts becomes small, a signal is present. The strong- est of two or more cochannel signals will be reflected in the subsequent counts for CIR > 7 dB. This phenomenon is the digital equivalent of FM capture [245, p. 497]. Random noise generates zero-crossings with large variance, but a sinusoid has a tight variance. Frequency modulations like GMSK exhibit two different zero-crossing rates, one corresponding to mark, and the other to space. The output of a zero-crossing counter, then, can be gated and reset at the expected channel-symbol rate. A threshold determines whether the channel symbol was mark or space, yielding the baseband bitstream. Timing logic can also estimate and track symbol timing. Although the logic has to operate at the GHz rates of the RF zero-crossings, the counter logic is simplicity itself.
9. RECEIVER ARCHITECTURES 273 Figure 8-5 Workable situation for roofing filter. Figure 8-6 Roofing filters distort subscriber signals. Not all situations can be addressed effectively using roofing filters, however. If there are more than a few strong interference signals in the passband, the roofing filters may introduce excessive distortion into the subscriber signals. This situation is illustrated in Figure 8-6. Factors that determine the number and characteristics of allowed roof- ing filters include the modulation of the subscriber signals, and the band- width of the interference relative to the overall passband. If the subscriber signals are robust to phase and amplitude distortion (e.g., FSK), then more filters or filters that introduce more severe distortion may be used. If the sub- scriber signals are phase-sensitive (e.g., 16 QAM proposed in many of the 3G alternatives), no more than one analog roofing filter is likely to be work- able. 3. Active Cancellation Active cancellation is the process of introducing a replica of the transmitted signal into the receiver so that it may be some-
10. 274 RF/IF CONVERSION SEGMENT TRADEOFFS how subtracted from the input signal. A detailed treatment of cancellation techniques is beyond the scope of this text, but the following introduces the essential notions. Active blanking of radar signals from the input to communications systems on the same platform is an example of active cancellation. In this case, the radar transmitter provides a control line that is active a few microseconds before it transmits so that the communications system can activate a grounding circuit. The RF stage passes no signal at all to the rest of the communications system until the control line is inactive [245]. Active communications cancellation circuits may delay the transmitted sig- nal and attenuate it in such a way that the transmitted and received signals are exactly out of phase, shifted by ¼ radians (at RF or IF) with respect to each other. In principle, such a circuit should cause the transmitted signal to be completely removed from the received signal. In practice, the cancellation is not ideal. In part, this is due to the inexactness of fabrication of analog circuits. In part, modulation of the transmitted signal distorts each IF sinusoid slightly, and the filtering-induced distortion through the transmitting antenna and into the receiving antenna (or through the circulator) differs slightly from the dis- tortion of the cancellation circuit. The result is that simple linear techniques can achieve only about 10 to 20 dB of cancellation. Complex phase-tracking circuits can improve performance, but nonlinear techniques are required to approach 30 to 40 dB. Few of the nonlinear techniques are in the public do- main. The cancellation that is needed is the difference between the maximum nondistorting input signal and the radiation level that reaches the receiving antenna. Required-Cancellation = (Peak energy at the output of the receiver antenna terminals) (Maximum linear energy) If this power is not suppressed or dissipated, it will capture the roof of the dynamic range and cause either intermodulation distortion or lost subscribers or both. Not all cancellation has to be accomplished using analog circuits. Any cancellation that occurs in the early stages of RF amplification and filtering also improves system linearity and contributes to dynamic range improvement just like roofing filters. Residual components may be further suppressed using digital techniques. 4. Software-based Interference Mitigation SDR architecture exacerbates interference mitigation by driving the radio platforms toward the use of wideband antennas and RF. It also can contribute to interference suppres-
11. RECEIVER ARCHITECTURES 275 TABLE 8-1 Mode Constraint Table (Minimal) Mode/ Constraint PTTj EPLRSj GSMj PTTi i, j < PTTmax; Pmax NPTT + NEPLRS < NPTT + NGSM < N max fPTTi " fPTTj < N max fPTT fPTT " fGSM < F min FPTT min "fEPLRS < F min EPLRSi RPTT + REPLRS < R max i, j < EPLRS max; NEPLRS + NGSM < P max N max GSMi RPTT + RGSM < R max RGSM + REPLRS < i, j < GSM max; R max P max PTT, EPLRS, i + j + k < N max Ri + Rj + Rk < Pi + Pj + Pk < GSM R max P max sion in several ways. A well-designed SDR has a table of constraints among combinations of waveforms that can operate simultaneously on the platform. The entries of the constraint table specify parametric limits on power, fre- quency, data rate, and number of simultaneous channels supported, as a min- imum. Table 8-1 provides a minimal example of a constraint table. In this case a notional dual-use military-commercial PDA has three possible waveforms: push-to-talk (PTT) AM/FM voice, EPLRS, and GSM. The entries on the di- agonal limit the number of channels that can be used in each mode to less than #mode$max, where #mode$ is PTT, EPLRS, or GSM. If the radio has four channels, it may be capable of supporting all four as push-to-talk chan- nels, but it may have some capacity limit to only one EPLRS channel and only two GSM channels. When used in combination, however, the number of PTT and GSM channels may not be the sum of the individual limits. The entries “N#mode1$+ N#mode2$ < N max” specify the limits when two modes are used in combination. In addition, the PTT row has been augmented with limits on the frequencies of the modes. The first column specifies that any two PTT channels must have the minimum frequency separation FPTT min. The other entries specify limits on the separation of combinations of modes. Additional entries specify joint limits on data rate (R#mode\$) when modes are used jointly. One may specify a total data rate for all subscribers that cannot be exceeded. Other constraints to be included in such a table are the presence and status of an active cancellation circuit, or the measured distance from the trans- mitting node to the nearest colocated node. This distance may be estimated using round-trip leading-edge delay techniques similar to the way radio dis- tance measuring equipment (DME) operates [399]. An SDR with a 100 MHz ADC/DAC channel and an FPGA with access to the digital IF signal could send a DME signal to be transponded by nearby radios. The internal delays can be calibrated so that the distance can be estimated to within 100 feet or so.